Microwave Engineering Europe - June 2008 - (Page 32)

32 RECEIVER ARCHITECTURE IP2 & IP3 design considerations with direct conversion I/Q demodulator receiver By Doug Stuetzle, Senior Design Engineer, Linear Technology Corp. Introduction Direct conversion receiver architecture offers several advantages over the traditional superheterodyne. It eases the requirements for RF front-end bandpass Þltering, as it is not susceptible to signals at the image frequency. The RF bandpass Þlters need only attenuate strong out-of-band signals to prevent them from overloading the front end. Also, direct conversion eliminates the need for IF ampliÞers and bandpass Þlters. Instead, the RF input signal is directly converted to baseband, where ampliÞcation and Þltering are much less difÞcult. The overall complexity and parts count of the receiver are reduced as well. Direct conversion does, however, come with its own set of implementation issues. Since the receive LO signal is at the same frequency as the RF signal, it can easily radiate from the receive antenna and violate regulatory standards. Unwanted baseband signals can also be generated by 2nd order nonlinearity of the receiver. A tone at any frequency entering the receiver will give rise to a DC offset in the baseband circuits. Once generated, straightforward elimination of DC offset becomes very problematic. That is because the frequency response of the post-downconversion circuits must often extend to DC. The 2nd order nonlinearity of the receiver also allows a modulated signal, even the desired signal, to generate a pseudo-random block of energy centered about DC. Unlike super-heterodyne receivers, direct conversion receivers are susceptible to such 2nd order mechanisms regardless of the frequency of the incoming signal. So minimizing the effect of Þnite 2nd order linearity is critical to the design of such receivers. Later in this paper we will consider the effect of 3rd order distortion on a direct conversion receiver. In this case, two signals separated by an appropriate frequency must enter the receiver in order for unwanted products to appear at the baseband frequencies. Second order distortion (IP2) The second order intercept point (IP2) of a direct conversion receiver system is a critical Figure 1: Block diagram of a typical WCDMA basestation receiver. performance parameter. It is a measure of second order non-linearity and helps quantify the receiverÕs susceptibility to single and two tone interfering signals. LetÕs examine how this nonlinearity affects sensitivity. We can model the transfer function of any nonlinear element as a Taylor series: y(t) = x(t) + a2x2 (t) + a3x3(t) + É, where x(t) is the input signal consisting of both desired and undesired signals. Consider only the second order distortion term for this analysis. The coefÞcient a2 is equal to sqrt(2/(ZoIP2)) where IP2 is the single tone intercept point in W. Note that the two-tone IP2 is 6 dB below the single-tone IP2. The more linear the element, the smaller a2 will be. Every signal entering the nonlinear element will generate a signal centered at zero frequency. Even the desired signal will give rise to distortion products at baseband. To illustrate this, let the input signal be represented by x(t) = A(t)cosωt, which may be a tone or a modulated signal. If it is a tone, then A(t) is simply a constant. If it is a modulated signal, then A(t) represents the signal envelope. By deÞnition, the power of the desired signal is 1/Zo * E{(A(t)cosωt)2}, where E{β} is the expected value of β. Since A(t) and cosωt are statistically independent, we can expand E{(A(t)cosωt)2} as E{A2(t)} * E{cos2ωt}. By trigonometry this is equal to E{A2(t)} * E{½+ ½cos2ωt}. The expected value of the second term is simply ½, so the resulting product is ½ * E{A2(t)}. The power of the desired signal simpliÞes to: Ps = 1/(2Zo) * E{A2(t)} (equation 1) In the case of a tone, A(t) may be replaced by A. The signal power is, as expected, equal to A2/(2Zo). In the more general case, the desired signal is digitally modulated by a pseudorandom data source. We can represent it as band-limited white noise with a Gaussian probability distribution. The signal envelope A(t) is now a Gaussian random variable. The expected value of the square of the envelope can be expressed in terms of the power of the desired signal as: E{A2(t)} = 2ZoPs (equation 2) Now substitute x(t) into the Taylor series expansion to Þnd y(t), which is the output of the nonlinear element: y(t) = A(t)cosωt + a2(A(t)cosωt)2 + É higher order terms = A(t)cosωt + ½a2A2(t) + ½a2A2(t) cos2ωt +É Consider the 2nd order distortion term ½a2A2(t). This term appears centered about DC, whereas the other 2nd order term appears near the 2nd harmonic of the desired signal. Only the term near DC is important here, as the high frequency tone will be rejected by the baseband circuitry. In the case where the signal is a tone, the 2nd order result is a DC offset equal to: DC offset = ½a2A2 = a2PsZo (equation 3) Microwave Engineering Europe ● June 2008 ● www.mwee.com http://www.mwee.com

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Microwave Engineering Europe - June 2008 Contents Comment News Cover Feature Designing and Simulating a Wireless LAN Antenna 60GHz: Achieving the Ultimate Wireless Dream New Radar Developments Include HFETs to Challenge DMOS/LDMOS and a 77-GHz CMOS PA for Automotive Applications Testing Raises Concerns Over 802.11-Based High-Speed Bluetooth IP2 & IP3 Design Considerations with Direct Conversion I/Q Demodulator Receiver Products Calendar

Microwave Engineering Europe - June 2008

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